The present invention relates to a preamplifier for use in optical communication receivers.
In general, a transimpedance-type preamplifier (FIG. 10(a)) and a high-impedance-type preamplifier (FIG. 10(b)) are employed in optical communication receivers. For example, these circuits are described in K. Ogawa, "Considerations for Optical Receiver Design," IEEE Journal on Selected Areas in Communications, Vol. SAC-1, No. 3, 1983, pp. 524-532.
Conventionally, repeaters used in a trunk system of a telephone network, which is the main application field of the optical communication, are required to be highly sensitive to amplify weak signals that have been subjected to attenuation through long-distance transmission while maintaining their waveforms.
On the other hand, in recent years, the application fields of the optical communication have expanded, for instance, to the data communication and subscriber systems. To be applicable to such a variety of communication systems, i.e., to accommodate various light sources, wide-range transmission distance, signal attenuation in optical fibers and communication network topology, etc., the development of an optical communication receiver is now desired which, in addition to being highly sensitive, can deal with optical signals having a wide dynamic range.
In the above circumstances, the present inventor proposed a preamplifier as shown in FIG. 11 (see, for instance, "Wide Dynamic Range GaAs Preamplifier IC for Lightwave Transmission," Autumn National Conference of the Institute of Electronics, Information and Communication Engineers, Presentation No. B-743, 1990). As shown in FIG. 11, a preamplifier A has the transimpedance-type basic constitution, and is incorporated in an optical receiver B that corresponds to the front-end portion of a repeater. The preamplifier A produces an output signal V.sub.OUT by amplifying an input signal V.sub.IN produced by impedance-conversion of a photocurrent that is an output of a photodetector PD receiving an optical signal h.upsilon. from a light transmission line. A bypass circuit C is provided to improve the dynamic range. That is, a cascade connection of a transimpedance-type phase-inverting amplifier 1 having a feedback resistor r.sub.f and an output buffer circuit 2 is provided between the input and output terminals of the preamplifier A. The source and drain of a field-effect transistor 3 of the bypass circuit C are connected to the respective terminals of the feedback resistor r.sub.f. A voltage in proportion to an output level of the output buffer circuit 2 is applied to the gate of the transistor 3 via a level shift circuit 4.
When the input signal V.sub.IN having an excessively large amplitude input signal is input to the preamplifier A, the field effect transistor 3 turns on in response to a variation of the output signal V.sub.OUT. Therefore, an effective feedback resistance R.sub.F (parallel resistance of the feedback resistance r.sub.f and the transistor 3) decreases to improve the dynamic range.
Referring to FIG. 12, a specific example of the transimpedance-type preamplifier A of FIG. 11 is described in detail. The preamplifier of FIG. 12 consists of compound semiconductor field-effect transistors (hereinafter referred to as FETs) T.sub.1 -T.sub.8 such as GaAs MESFETs, level-shift diode groups d.sub.1 and d.sub.2, resistors r.sub.1 and r.sub.2 and a feedback resistor r.sub.f, and operates on a single supply voltage V.sub.DD. The phase-inverting preamplifier 1 is formed by the FETs T.sub.1 -T.sub.4 and the diode groups d.sub.1 and d.sub.2, the output buffer circuit 2 is formed by the FETs T.sub.5 and T.sub.6, and the level shift circuit 4 is formed by the FET T.sub.7 and the resistors r.sub.1 and r.sub.2. The feedback resistor r.sub.f and the FET T.sub.8 in FIG. 12 correspond to the resistor r.sub.f and the FET 3 in FIG. 10, respectively. The threshold voltage V.sub.T8 of the FET T.sub.8 is -0.5 V, and the bias setting is so made that the gate-source voltage V.sub.GS of the FET T.sub.8 is lower than -0.5 V and no current flows between the source and drain when no signal or a very small signal (e.g., smaller than -20 dBm) is input.
As the amplitude of the input signal V.sub.IN increases in response to the rise of the input optical signal intensity, the voltage level of the output signal V.sub.OUT decreases as a result of the phase-inverting amplification and both of the gate voltage V.sub.G and the source voltage V.sub.S of the FET T.sub.8 decrease. Since the source voltage V.sub.s drops more than the gate voltage V.sub.G, the gate-source voltage V.sub.GS increases. As a result, when the input optical signal intensity exceeds a certain value (e.g., -10 dBm), the FET T.sub.8 turns on to reduce the transimpedance. Even if the amplitude of the input signal V.sub.IN is further increased, the reduction of the transimpedance causes clipping of voltage variations within the preamplifier. Therefore, the respective FETs T.sub.1 -T.sub.7 are not so biased as to work in the non-saturation region, which means the increase of the maximum allowable input level (maximum allowable amplitude of the input signal V.sub.IN). That is, the preamplifier of FIGS. 11 and 12 can increase the dynamic range by raising the maximum allowable input level.
On the other hand, in the preamplifier described above, the feedback resistor r.sub.f needs to have a large value to lower the minimum sensible light intensity within the dynamic range. That is, since the thermal noise &lt;i.sub.RF.sup.2 &gt; decreases as the feedback resistance r.sub.f increases (see equation (1) below), the minimum sensible light intensity can be lowered. ##EQU1## where R.sub.F is the effective feedback resistance between the input and output of the amplifier 1, W is a frequency bandwidth of the preamplifier, T is the temperature and k is the Boltzmann constant.
However, as is seen from equation (2) belows simply increasing the effective feedback resistance R.sub.F causes a problem that a frequency bandwidth .omega..sub.C of the optical receiver B is reduced. ##EQU2## where G is an open-loop gain of the amplifier 1, C.sub.T is an input capacitance, and R.sub.F is the effective feedback resistance.
Therefore, as is understood from equation (2), the open-loop gain G of the amplifier 1 needs to be increased or the input capacitance needs to be decreased to provide the large effective resistance R.sub.F (for the improvement of the minimum sensible light intensity) and secure the sufficiently wide bandwidth .omega..sub.C of the optical receiver B.
The present inventor proposed a preamplifier shown in FIG. 13 which can increase the open-loop gain G (see, for instance, "High Gain and Broadband GaAs Preamplifier IC's for High Speed Optical Receivers," Autumn National Conference of the Institute of Electronics, Information and Communication Engineers, Presentation No. B-744, 1990). The open-loop gain is increased by employing a current-injection-type circuit using FETs having two different threshold voltages. The preamplifier of FIG. 13 has the basic constitution in which a FET T.sub.IN for current injection is provided between the drain of the input-stage FET T.sub.1 and the supply voltage V.sub.DD of the preamplifier of FIG. 12, and a FET T.sub.IS is added to isolate the FET T.sub.IN from the FET T.sub.2 serving as a load. The feedback resistor r.sub.f is provided between the input and output terminals.
In the preamplifier of FIG. 13, the open-loop gain G is increased by current injection from the current injection FET T.sub.IN to the input-stage FET T.sub.1, to a large value of about 33 dB. However, according to the general principle of constant gain-bandwidth product, the bandwidth .omega..sub.C decreases, to about 600 MHz, compared to the case of not incorporating the measure for increasing the gain G (e.g., the preamplifier of FIG. 12).
To cope with this problem, a preamplifier of FIG. 14 was developed by combining the advantages of the first conventional preamplifier of FIGS. 11 and 12 which can increase the maximum allowable input level by incorporation of the bypass circuit C and the second conventional preamplifier of FIG. 13 which can lower the minimum sensible light intensity.
However, this simple combination could not provide an optimum preamplifier because the following problems actually occurred. In the preamplifier of FIG. 14, the bypass circuit is formed by adding a switching diode d.sub.S (corresponding to the FET T.sub.8 in FIG. 12) to the preamplifier of FIG. 13 having the current injection FET T.sub.IN. If the frequency characteristic of the preamplifier has a first-order pole, a transimpedance transfer function Z.sub.T (s) of an optical receiver including it has a second-order pole and is therefore expressed by equation (3) where .omega..sub.C is an angular frequency bandwidth of the optical receiver, G.sub.O is a d.c. open-loop gain of the preamplifier, .omega..sub.h is an angular frequency bandwidth of the preamplifier, R.sub.F is an effective feedback resistance (i.e., a parallel resistance of the feedback resistor r.sub.f and the switching diode d.sub.S), and C.sub.T is an input capacitance of the preamplifier. ##EQU3##
The problems occur in the following manner. When an excessively large input signal V.sub.IN comes in, the bypass circuit operates to reduce the effective feedback resistance R.sub.F, which means decrease of .zeta. in equation (3). Since the decrease of .zeta. means increase of the feedback quantity, the bandwidth .omega..sub.C of the optical receiver increases to become close to the upper limit of the bandwidth .omega..sub.h of the preamplifier as the input signal V.sub.IN further increases to further reduce .zeta., as seen from equation (2). In this situation, a condition .omega..sub.C .gtoreq..omega..sub.h is more likely to be established than in the first conventional case, because, as described above, the bandwidth .omega..sub.h of the preamplifier whose gain is increased by the current injection is narrower than the first conventional preamplifier of FIG. 12. As a result, as is understood from equation (3), the optical receiver exhibits a peaking-type transimpedance characteristic as shown in FIG. 15(a) and its operation becomes unstable.
With the peaking-type characteristic, a rectangular NRZ (non-return-to-zero) optical signal will cause ringing or oscillation (FIG. 15(b)), so that the output signal V.sub.OUT will not assume a waveform faithful to the input signal waveform. In an actual measurement (see FIG. 16), an output signal V.sub.OUT having a waveform distortion associated with an oscillation phenomenon was observed in response to a NRZ input optical signal having an average light intensity of -10 dBm and a pulse rate of 622 Mbit/sec.
In summary, when the input signal V.sub.IN is not too large, the bandwidth .omega..sub.h of the preamplifier is sufficiently larger than the bandwidth .omega..sub.C of the optical receiver (.omega..sub.h &gt;.omega..sub.C) and the stable operation is obtained. However, when the amplitude of the input signal V.sub.IN becomes too large, .omega..sub.C increases to become close to .omega..sub.h or to exceed it, in which case the optical receiver becomes unstable and exhibits, for instance, an oscillation.